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FEATURES
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LTC3606B 800mA Synchronous Step-Down DC/DC with Average Input Current Limit DESCRIPTION
The LTC(R)3606B is an 800mA monolithic synchronous buck regulator using a constant frequency current mode architecture. The input supply voltage range is 2.5V to 5.5V, making it ideal for Li-Ion and USB powered applications. 100% duty cycle capability provides low dropout operation, extending the run time in battery-operated systems. Low output voltages are supported with the 0.6V feedback reference voltage. The LTC3606B can supply 800mA output current. The LTC3606B's programmable average input current limit is ideal for USB applications and for point-of-load power supplies because the LTC3606B's limited input current will still allow its output to deliver high peak load currents without collapsing the input supply. The operating frequency is internally set at 2.25MHz allowing the use of small surface mount inductors. Internal soft-start reduces in-rush current during start-up. The LTC3606B is available in an 8-Lead 3mm x 3mm DFN package.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815, 6304066, 6498466, 6580258, 6611131.
Programmable Average Input Current Limit: 5% Accuracy Step-Down Output: Up to 96% Efficiency Low Noise Pulse-Skipping Operation at Light Loads Input Voltage Range: 2.5V to 5.5V Output Voltage Range: 0.6V to 5V 2.25MHz Constant-Frequency Operation Power Good Output Voltage Monitor Low Dropout Operation: 100% Duty Cycle Internal Soft-Start Current Mode Operation for Excellent Line and Load Transient Response 2% Output Voltage Accuracy Short-Circuit Protected Shutdown Current 1A Available in Small Thermally Enhanced 8-Lead 3mm x 3mm DFN Package
APPLICATIONS
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High Peak Load Current Applications USB Powered Devices Supercapacitor Charging Radio Transmitters and Other Handheld Devices
TYPICAL APPLICATION
Monolithic Buck Regulator with Input Current Limit
VIN 3.4V TO 5.5V CIN 10F PGOOD 499k 1.5H VIN LTC3606B RUN PGOOD RLIM GND 1000pF 116k ILIM = 475mA VFB 1210k SW VOUT 3.4V AT 800mA VOUT 200mV/DIV VIN AC-COUPLED 1V/DIV IOUT 500mA/DIV
3606B TA01
GSM Pulse Load
+
2.2mF 2 SuperCap
255k
IIN 500mA/DIV 1ms/DIV VIN = 5V, 500mA COMPLIANT ILOAD = 0A to 2.2A
3606B TA01b
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LTC3606B ABSOLUTE MAXIMUM RATINGS
(Note 1)
PIN CONFIGURATION
TOP VIEW 1 2 3 4 9 GND 8 VFB 7 RUN 6 PGOOD 5 VIN GND RLIM GND SW
Input Supply Voltage (VIN) ........................... -0.3V to 6V VFB ................................................... -0.3V to VIN + 0.3V RUN, RLIM ....................................... -0.3V to VIN + 0.3V SW ................................................... -0.3V to VIN + 0.3V PGOOD............................................. -0.3V to VIN + 0.3V P-Channel SW Source Current (DC) (Note 2) ..............1A N-Channel SW Source Current (DC) (Note 2) .............1A Peak SW Source and Sink Current (Note 2) ............. 2.7A Operating Junction Temperature Range (Notes 3, 6, 8) ........................................ -40C to 125C Storage Temperature Range .................. -65C to 125C Reflow Peak Body Temperature ............................ 260C
DD PACKAGE 8-LEAD (3mm 3mm) PLASTIC DFN TJMAX = 125C, JA = 40C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH LTC3606BEDD#PBF LTC3606BIDD#PBF TAPE AND REEL LTC3606BEDD#TRPBF LTC3606BIDD#TRPBF PART MARKING* LFMB LFMB PACKAGE DESCRIPTION 8-Lead (3mm x 3mm) Plastic DFN 8-Lead (3mm x 3mm) Plastic DFN TEMPERATURE RANGE -40C to 85C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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LTC3606B ELECTRICAL CHARACTERISTICS
SYMBOL VIN VUV IFB VFBREG VLINEREG VLOADREG IS PARAMETER VIN Operating Voltage Range VIN Undervoltage Lockout Feedback Pin Input Current Feedback Voltage VFB Line Regulation VFB Load Regulation Supply Current Active Mode (Note 4) Shutdown Oscillator Frequency Peak Switch Current Limit Input Average Current Limit LTC3606BE, -40C < TJ < 85C (Note 7) LTC3606BI, -40C < TJ < 125C (Note 7) VIN = 2.5V to 5.5V (Note 7) ILOAD = 0mA to 800mA (Note 7) VFB = 0.95 x VFBREG VRUN = 0V, VIN = 5.5V VFB = VFBREG VIN = 5V, VFB < VFBREG , Duty Cycle <35% RLIM = 116k RLIM = 116k, LTC3606BE RLIM = 116k, LTC3606BI
l l l
The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25C, VIN = 5V, unless otherwise noted.
CONDITIONS
l
MIN 2.5
l l l l
TYP 2.1
MAX 5.5 2.5 30 0.612 0.618 0.25
UNITS V V nA V V %/V %
VIN Low to High
0.588 0.582
0.600 0.600 0.01 0.5 420
650 1 2.7 500 513 523
A A MHz mA mA mA mA
fOSC ILIM(PEAK) IINLIM
1.8 1800 450 437 427
2.25 2400 475 475 475 0.27 0.25 0.01
RDS(ON) ISW(LKG) tSOFTSTART VRUN IRUN PGOOD
Main Switch On-Resistance (Note 5) VIN = 5V, ISW = 100mA Synchronous Switch On-Resistance (Note 5) VIN = 5V, ISW = 100mA Switch Leakage Current Soft-Start Time RUN Threshold High RUN Leakage Current Power Good Threshold 0V VRUN 5V Entering Window VFB Ramping Up VFB Ramping Down Leaving Window VFB Ramping Up VFB Ramping Down PGOOD Rising and Falling, VIN = 5V 8 VPGOOD = 5V VIN = 5V, VRUN = 0V VFB from 0.06V to 0.54V
l l
1 1.3 1.2 1
A ms V A % %
0.3 0.4
0.95 1 0.01
-5 5
-7 7 9 -9 90 15 30 1 11 -11
% % s A
PGOOD Blanking Power Good Blanking Time RPGOOD IPGOOD Power Good Pull-Down On-Resistance PGOOD Leakage Current
Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Guaranteed by long term current density limitations. Note 3: The LTC3606BE is guaranteed to meet performance specifications from 0C to 85C. Specifications over the -40C to 85C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3606BI is guaranteed to meet specified performance over the full -40C to 125C operating junction temperature range. Note 4: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency.
Note 5: The switch on-resistance is guaranteed by correlation to wafer level measurements. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 7: The converter is tested in a proprietary test mode that connects the output of the error amplifier to the SW pin, which is connected to an external servo loop. Note 8: TJ is calculated from the ambient temperature TA and the power dissipation as follows: TJ = TA + (PD)(JAC/W)
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LTC3606B TYPICAL PERFORMANCE CHARACTERISTICS
Pulse-Skipping Mode Operation
550 SW 2V/DIV SUPPLY CURRENT (A) VOUT 50mV/DIV ACCOUPLED 500 450 EFFICIENCY (%) 400 350 300 250 200 -50 VIN = 2.7V
TA = 25C, VIN = 5V, unless otherwise noted.
Supply Current vs Temperature
RUN = VIN ILOAD = 0A VIN = 5.5V 100 90 80 70 60 50 40 30 20
Efficiency vs Input Voltage
IL 100mA/DIV VIN = 5V VOUT = 3.3V ILOAD = 5mA 5s/DIV
3606B G01
IOUT = 100mA IOUT = 400mA IOUT = 800mA VOUT = 3.3V 4 4.5 VIN (V)
IOUT = 10mA IOUT = 1mA IOUT = 0.1mA
10 -25 0 25 50 75 TEMPERATURE (C) 100 125
0 3.5
5
5.5
3606B G03
3606B G02
Regulated Voltage vs Temperature
1.5 1.0 VFB ERROR (%) 0.5 0 -0.5 -1.0 -1.5 -50 FREQUENCY (MHz) 2.5 2.4
Oscillator Frequency vs Temperature
1000
Switch Leakage vs Input Voltage
800 2.3 2.2 2.1 2.0 1.9 1.8 -50 LEAKAGE CURRENT (pA)
600 MAIN SWITCH 400 SYNCHRONOUS SWITCH
VIN = 2.7V VIN = 3.6V VIN = 4.2V VIN = 5V -25 0 25 50 75 TEMPERATURE (C) 100 125
200
-25
0 25 50 75 TEMPERATURE (C)
100
125
0 2.5
3
3.5
3606B G04
4 VIN (V)
4.5
5
5.5
3606B G06
3606B G05
Switch On-Resistance vs Input Voltage
600 0.5 0.4 MAIN PFET RDS(ON) () 500 RDS(ON) (m) 0.3 0.2 0.1 0 SYNCHRONOUS SWITCH 200 2.5 3 3.5 4 VIN (V) 4.5 5 5.5
3606B G07
Switch On-Resistance vs Temperature
VIN = 2.7V VIN = 3.6V VIN = 5V 0.7 MAIN SWITCH 0.6 0.5 0.4 0.3 0.2 SYNCHRONOUS SWITCH -0.1 -50 -25 25 50 75 0 TEMPERATURE (C) 100 0.1 125 SYNCHRONOUS NFET RDS(ON) () 100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10
Efficiency vs Load Current
VOUT = 3.3V
400 MAIN SWITCH 300
0 0.0001
VIN = 3.6V VIN = 4.2V VIN = 5V 0.001 0.01 0.1 OUTPUT CURRENT (A) 1
3606B G11
3606B G09
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LTC3606B TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current
100 90 80 VOUT ERROR (%) VOUT ERROR (%) EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.0001 VIN = 2.7V VIN = 3.6V VIN = 4.2V VIN = 5V 0.001 0.01 0.1 OUTPUT CURRENT (A) 1
3606B G13
TA = 25C, VIN = 5V, unless otherwise noted.
Load Regulation
3.0 2.5 2.0 0.2 0 -0.2 -0.4 1.5 1.0 0.5 0 -0.5 -1.0 0 VOUT = 1.8V VOUT = 2.5V VOUT = 3.3V 100 200 300 400 500 600 700 800 LOAD CURRENT (mA)
3606B G15
Line Regulation
0.6 0.4 VOUT = 1.8V ILOAD = 100mA
VOUT = 1.2V
-0.6 2.5
3.0
3.5
4.0 VIN (V)
4.5
5.0
5.5
3606B G16
Start-Up from Shutdown
RUN 2V/DIV RUN 2V/DIV VOUT 2V/DIV VOUT 1V/DIV IL 250mA/DIV 200s/DIV VIN = 5V, VOUT = 3.3V RLOAD = 7 CLOAD = 4.7F
3606B G17
Start-Up from Shutdown
1.2 1.0 0.8 VRLIM (V) 0.6 0.4 0.2 0
VRLIM vs Input Current
ILIM = 475mA RLIM = 116k
RLIM 1V/DIV IIN 500mA/DIV 2ms/DIV VIN = 5V, VOUT = 3.4V RL = NO LOAD, CL = 4.4mF , CLIM = 2200pF ILIM = 500mA
3606B G18
0
100
200
300 400 IIN (mA)
500
600
3606B G18b
Average Input Current Limit vs Temperature
8 VIN = 5V 6 ILIM = 475mA 4 IINLIM ERROR (%) 2 0 -2 -4 -6 -8 -50 -25 0 25 50 75 TEMPERATURE (C) 100 125 IL 1A/DIV ILOAD 1A/DIV VOUT 200mV/DIV AC-COUPLED
Load Step
VOUT 200mV/DIV AC-COUPLED
Load Step
IL 1A/DIV ILOAD 1A/DIV 20s/DIV VIN = 5V, VOUT = 3.3V ILOAD = 0A TO 800mA , COUT = 100F CF = 20pF
3606B G20
20s/DIV VIN = 5V, VOUT = 1.8V ILOAD = 80mA TO 800mA , COUT = 100F CF = 20pF
3606B G21
3606B G19
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LTC3606B PIN FUNCTIONS
GND (Pins 1, 3, Exposed Pad Pin 9): Ground. Connect to the (-) terminal of COUT, and the (-) terminal of CIN. The Exposed Pad must be soldered to PCB. RLIM (Pin 2): Average Input Current Limit Program Pin. Place a resistor and capacitor in parallel from this pin to ground. SW (Pin 4): Regulator Switch Node Connection to the Inductor. This pin swings from VIN to GND. VIN (Pin 5): Main Power Supply. Must be closely decoupled to GND. PGOOD (Pin 6): Open-Drain Logic Output. PGOOD is pulled to ground if the voltage on the VFB pin is not within power good threshold. RUN (Pin 7): Regulator Enable. Forcing this pin to VIN enables regulator, while forcing it to GND causes regulator to shut down. VFB (Pin 8): Regulator Output Feedback. Receives the feedback voltage from the external resistive divider across the regulator output. Nominal voltage for this pin is 0.6V.
FUNCTIONAL DIAGRAM
RUN 7 0.6V REF OSC
OSC
+ -
2 RLIM 1V
MIN CLAMP SLOPE COMP 5 VIN
VFB
8 0.6V
- - +
EA
ITH
-
VSLEEP
SLEEP
- +
S Q RS LATCH R Q SWITCHING LOGIC AND BLANKING CIRCUIT ICOMP
+
SOFT-START
+
ICOMP
-
ANTI SHOOTTHRU 4 SW
IRCMP SHUTDOWN
6
-
+
+ -
PGOOD
6
+ -
0.654V VFB 0.546V
9 GND
3606B FD
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LTC3606B OPERATION
The LTC3606B uses a constant-frequency, current mode architecture. The operating frequency is set at 2.25MHz. The output voltage is set by an external resistor divider returned to the VFB pins. An error amplifier compares the divided output voltage with a reference voltage of 0.6V and regulates the peak inductor current accordingly. The LTC3606B continuously monitors the input current via the voltage drop across the RDS(ON) of the internal P-channel MOSFET. When the input current exceeds the programmed input current limit set by an external resistor, RLIM , the regulator's input current is limited. The regulator output voltage will drop to meet output current demand and to maintain constant input current. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle when the VFB voltage is below the reference voltage. The current into the inductor and the load increases until the peak inductor current (controlled by ITH) is reached. The RS latch turns off the synchronous switch and energy stored in the inductor is discharged through the bottom switch (N-channel MOSFET) into the load until the next clock cycle begins, or until the inductor current begins to reverse (sensed by the IRCMP comparator). The peak inductor current is controlled by the internally compensated ITH voltage, which is the output of the error amplifier. This amplifier regulates the VFB pin to the internal 0.6V reference by adjusting the peak inductor current accordingly. When the input current limit is engaged, the peak inductor current will be lowered, which then reduces the switching duty cycle and VOUT. This allows the input voltage to stay regulated when its programmed current output capability is met. Light Load Operation The LTC3606B will automatically transition from continuous operation to the pulse-skipping operation when the load current is low. The inductor current is not fixed during the pulse-skipping mode which allows the LTC3606B to switch at constant-frequency down to very low currents, where it will begin skipping pulses to maintain output regulation. This mode of operation exhibits low output ripple as well as low audio noise and reduced RF interference while providing reasonable low current efficiency. Dropout Operation When the input supply voltage decreases toward the output voltage the duty cycle increases to 100%, which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. An important design consideration is that the RDS(ON) of the P-channel switch increases with decreasing input supply voltage (see the Typical Performance Characteristics section). Therefore, the user should calculate the worstcase power dissipation when the LTC3606B is used at 100% duty cycle with low input voltage (see Thermal Considerations in the Applications Information section). Soft-Start In order to minimize the inrush current on the input bypass capacitor, the LTC3606B slowly ramps up the output voltage during start-up. Whenever the RUN pin is pulled high, the corresponding output will ramp from zero to full-scale over a time period of approximately 750s. This prevents the LTC3606B from having to quickly charge the output capacitor and thus supplying an excessive amount of instantaneous current. When the output is loaded heavily, for example, with millifarad of capacitance, it may take longer than 750s to charge the output to regulation. If the output is still low after the soft-start time, the LTC3606B will try to quickly charge the output capacitor. In this case, the input current limit (after it engages) can prevent excessive amount of instantaneous current that is required to quickly charge the output. See the Start-Up from Shutdown curve (CL = 4.4mF)in the Typical Performance Characteristics section. After input current limit is engaged, the output slowly ramps up to regulation while limited by its 500mA of input current.
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LTC3606B OPERATION
Short-Circuit Protection When either regulator output is shorted to ground, the corresponding internal N-channel switch is forced on for a longer time period for each cycle in order to allow the inductor to discharge, thus preventing inductor current runaway. This technique has the effect of decreasing switching frequency. Once the short is removed, normal operation resumes and the regulator output will return to its nominal voltage. Input Current Limit Internal current sense circuitry measures the inductor current through the voltage drop across the power PFET switch and forces the same voltage across the small sense PFET. The voltage across the small sense PFET generates a current representing 1/55,000th of the inductor current during the on-cycle. The current out of RLIM pin is the representation of the inductor current, which can be expressed in the following equation. IRLIM = IOUT * D1 * K1 where D1 = VOUT1/VIN is the duty cycle. K1 is the ratio RDS(ON) (power PFET)/RDS(ON)(sense PFET). The ratio of the power PFET to the sense PFET is trimmed to within 2%. Given that both PFETs are carefully laid out and matched, their temperature and voltage coefficient effects will be similar and their terms be canceled out in the equation. In that case, the constant K1 will only be dependent on area scaling, which is trimmed to within 2%. Thus, the IRLIM current will track the input current very well over varying temperature and VIN. The RLIM pin can be grounded to disable input current limit function. Programming Input Current Limit Selection of one external RLIM resistor will program the input current limit. The current limit can be programmed from 200mA up to IPEAK current. As the input current increases, RLIM voltage will follow. When RLIM reaches the internal comparator threshold of 1V, the power PFET on-time will be shortened, thereby, limiting the input current. Use the following equation to select the RLIM resistance that corresponds to the input current limit. RLIM = 55k / IDC IDC is the input current (at VIN) to be limited. The following are some RLIM values with the corresponding current limit.
RLIM 91.6k 110k 137.5k IDC 600mA 500mA 400mA
Selection of CLIM Capacitance Since IRLIM current is a function of the inductor current, its dependency on the duty cycle cannot be ignored. Thus, a CLIM capacitor is needed to integrate the IRLIM current and smooth out transient currents. The LTC3606B is stable with any size capacitance >100pF at the RLIM pin. Each application input current limit will call for different CLIM value to optimize its response time. Using a large CLIM capacitor requires longer time for the RLIM pin voltage to charge. For example, consider the application 500mA input current limit, 5V input and 1A, 2.5V output with a 50% duty cycle. When an instantaneous 1A output pulse is applied, the current out of the RLIM pin becomes 1A/55k = 18.2A during the 50% on-time or 9.1A full duty cycle. With a CLIM capacitor of 1F, RLIM of 116k, and using I = CdV/dt, it will take 110ms for CLIM to charge from 0V to 1V. This is the time after which the LTC3606B will start input current limiting. Any current within this time must be considered in each application to determine if it is tolerable.
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LTC3606B OPERATION
Figure 1a shows VIN (IIN) current below input current limit with a CLIM capacitor of 0.1F. When the load pulse is applied, under the specified condition, ILIM current is 1.1A/55k * 0.66 = 13.2A, where 0.66 is the duty cycle. It will take a little more than 7.5ms to charge the CLIM capacitor from 0V to 1V, after which the LTC3606B begins to limit input current. The IIN current is not limited during this 7.5ms time and is more than 725mA. This current transient may cause the input supply to temporarily droop if the supply current compliance is exceeded, but recovers after the input current limit engages. The output will continue to deliver the required current load while the output voltage droops to allow the input voltage to remain regulated during input current limit. For applications with short load pulse duration, a smaller CLIM capacitor may be the better choice as in the example shown in Figure 1b. In this example, a 577s, 0A to 2A output pulse is applied once every 4.7ms. A CLIM capacitor of 2.2nF requires 92s for VRLIM to charge from 0V to 1V. During this 92s, the input current limit is not yet engaged
VOUT 2V/DIV IIN 500mA/DIV VRLIM 1V/DIV IL 1A/DIV 50ms/DIV
3606B F01a
and the output must deliver the required current load. This may cause the input voltage to droop if the current compliance is exceeded. Depending on how long this time is, the VIN supply decoupling capacitor can provide some of this current before VIN droops too much. In applications with a bigger VIN supply decoupling capacitor and where VIN supply is allow to droop closer to dropout, the CLIM capacitor can be increased slightly. This will delay the start of input current limit and artificially regulated VOUT before input current limit is engaged. In this case, within the 577s load pulse, the VOUT voltage will stay artificially regulated for 92s out of the total 577s before the input current limit activates. This approach may be used if a faster recovery on the output is desired. Selecting a very small CLIM will speed up response time but it can put the device within threshold of interfering with normal operation and input current limit in every few switching cycles. This may be undesirable in terms of noise. Use 2RC >> 100/clock frequency (2.25MHz) as a starting point, R being RLIM, C being CLIM.
VOUT 200mV/DIV VIN AC-COUPLED 1V/DIV
IOUT 500mA/DIV IIN 500mA/DIV 1ms/DIV
3606B F01b
VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 0.1F , ILOAD = 0A to 1.1A, COUT = 2.2mF VOUT = 3.3V ILIM = 475mA
VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 2200pF , ILOAD = 0A to 2A, COUT = 2.2mF VOUT = 3.3V ILIM = 475mA
Figure 1a. Input Current Limit Within 100ms Load Pulses
Figure 1b. Input Current Limit Within 577s, 2A Repeating Load Pulses
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LTC3606B APPLICATIONS INFORMATION
A general LTC3606B application circuit is shown in Figure 2. External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected. Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current IL decreases with higher inductance and increases with higher VIN or VOUT : V V IL = OUT * 1 OUT (1) fO * L VIN Accepting larger values of IL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is 40% of the maximum output load current. So, for a 800mA regulator, IL = 320mA (40% of 800mA). The inductor value will also have an effect on Burst Mode operation. The transition to low current operation begins when the peak inductor current falls below a level set by the internal burst clamp. Lower inductor values result in higher ripple current which causes the transition to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. Furthermore, lower inductance values will cause the bursts to occur with increased frequency.
VIN 2.5V TO 5.5V RPGD CIN L1 VIN LTC3606B RUN PGOOD PGOOD RLIM GND RLIM CLIM
3606B F02
Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price versus size requirements, and any radiated field/EMI requirements, than on what the LTC3606B requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3606B applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER Coilcraft PART NUMBER LPS4012-152ML LPS4012-222ML LPS4012-332ML LPS4012-472ML LPS4018-222ML LPS4018-332ML LPS4018-472ML FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D ELT5KT4R7M CDRH2D18/LD CDH38D11SNP3R3M CDH38D11SNP2R2M MAX DC VALUE CURRENT 1.5H 2.2H 3.3H 4.7H 2.2H 3.3H 4.7H 4.7H 3.3H 2.2H 4.7H 4.7H 3.3H 2.2H 2.2H 2.2H 3.3H 2.2H 4.7H 4.7H 3.3H 2.2H 2.2H 3.3H 2.2H 2200mA 1750mA 1450mA 1450mA 2300mA 2000mA 1800mA 1100mA 1200mA 1300mA 450mA 950mA 630mA 1560mA 1900mA 510mA 530mA 410mA 1100mA 750mA 700mA 870mA 1000mA 1500mA 1700mA 2300mA DCR 0.070 0.100 0.100 0.170 0.070 0.080 0.125 0.11 0.1 0.08 0.2 0.2 0.086 0.115 0.082 0.13 0.33 0.27 0.1 0.19 0.28 0.17 0.12 0.076 0.095 0.059 HEIGHT 1.2mm 1.2mm 1.2mm 1.2mm 1.8mm 1.8mm 1.8mm 1mm 1mm 1mm 2mm 1.2mm 2mm 1.2mm 1.2mm 1.6mm 1.25mm 1.6mm 1mm 1mm 1mm 1mm 1mm 1.2mm 1.2mm 1.4mm
FDK
Murata Panasonic Sumida
LQH32CN4R7M23 4.7H
SW CF
VOUT COUT
Taiyo Yuden CB2016T2R2M CB2012T2R2M CB2016T3R3M NR30102R2M NR30104R7M TDK VLF3010AT4R7MR70 VLF3010AT3R3MR87 VLF3010AT2R2M1R0 VLF4012AT-2R2 M1R5 VLF5012ST-3R3 M1R7 VLF5014ST-2R2 M2R3
VFB
R2
R1
Figure 2. LTC3606B General Schematic
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LTC3606B APPLICATIONS INFORMATION
Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT / VIN . To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT (VIN VOUT ) VIN voltage, the output ripple is highest at maximum input voltage since IL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP , Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3606B control loop does not depend on the output capacitor's ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. For more information, see Application Note 88. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size.
Where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM - IL /2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer's ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1F to 1F ceramic capacitor is also recommended on VIN for high frequency decoupling when not using an all-ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple VOUT is determined by: VOUT 1 IL ESR+ 8fOCOUT
where fO = operating frequency, COUT = output capacitance and IL = ripple current in the inductor. For a fixed output
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LTC3606B APPLICATIONS INFORMATION
Setting the Output Voltage The LTC3606B regulates the VFB pin to 0.6V during regulation. Thus, the output voltage is set by a resistive divider, Figure 2, according to the following formula: VOUT = 0.6V 1+ R2 R1 (2) The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. In some applications, a more severe transient can be caused by switching in loads with large (>1F) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot SwapTM controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting. Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: % Efficiency = 100% - (L1 + L2 + L3 + ...) where L1, L2, etc., are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four sources usually account for the losses in LTC3606B circuits: 1) VIN quiescent current, 2) switching losses, 3) I2R losses, 4) other system losses. 1. The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load.
Keeping the current small (< 10A) in these resistors maximizes efficiency, but making it too small may allow stray capacitance to cause noise problems or reduce the phase margin of the error amp loop. To improve the frequency response of the main control loop, a feedback capacitor (CF) may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD * ESR, where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine the phase margin. In addition, feedback capacitors (CF) can be added to improve the high frequency response, as shown in Figure 2. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin.
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LTC3606B APPLICATIONS INFORMATION
2. The switching current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3. I2R losses are calculated from the DC resistances of the internal switches, RSW , and external inductor, RL. In continuous mode, the average output current flows through inductor L, but is "chopped" between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: RSW = (RDS(ON)TOP) * (DC) + (RDS(ON)BOT) * (1- DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT 2 * (RSW + RL) 4. Other "hidden" losses, such as copper trace and internal battery resistances, can account for additional efficiency degradations in portable systems. It is very important to include these "system" level losses in the design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other losses, including diode conduction losses during dead-time, and inductor core losses, generally account for less than 2% total additional loss. Thermal Considerations In a majority of applications, the LTC3606B does not dissipate much heat due to its high efficiency. In the unlikely event that the junction temperature somehow reaches approximately 150C, both power switches will be turned off and the SW node will become high impedance. The goal of the following thermal analysis is to determine whether the power dissipated causes enough temperature rise to exceed the maximum junction temperature (125C) of the part. The temperature rise is given by: TRISE = PD * JA where PD is the power dissipated by the regulator and JA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: TJ = TRISE + TAMBIENT As a worst-case example, consider the case when the LTC3606B is in dropout at an input voltage of 2.7V with a load current of 800mA and an ambient temperature of 70C. From the Typical Performance Characteristics graph of Switch Resistance, the RDS(ON) of the switch is 0.33. Therefore, the power dissipated is: PD = IOUT 2 * RDS(ON) = 212mV Given that the thermal resistance of a properly soldered DFN package is approximately 40C/W, the junction temperature of an LTC3606B device operating in a 70C ambient temperature is approximately: TJ = (0.212W * 40C/W) + 70C = 78.5C which is well below the absolute maximum junction temperature of 125C.
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LTC3606B APPLICATIONS INFORMATION
PC Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3606B. These items are also illustrated graphically in the layout diagrams of Figures 3a and 3b. Check the following in your layout: 1. Does the capacitor CIN connect to the power VIN (Pin 5) and GND (Pin 9) as closely as possible? This capacitor provides the AC current of the internal power MOSFETs and their drivers. 2. Are the respective COUT and L closely connected? The (-) plate of COUT returns current to GND and the (-) plate of CIN. 3. The resistor divider, R1 and R2, must be connected between the (+) plate of COUT and a ground sense line terminated near GND (Pin 9). The feedback signal VFB should be routed away from noisy components and traces, such as the SW line (Pin 4), and their trace length should be minimized. 4. Keep sensitive components away from the SW pin, if possible. The input capacitor CIN, CLIM and the resistors R1, R2, and RLIM should be routed away from the SW traces and the inductors. 5. A ground plane is preferred, but if not available, keep the signal and power grounds segregated with small signal components returning to the GND pin at a single point. These ground traces should not share the high current path of CIN or COUT. 6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. These copper areas should be connected to VIN or GND.
VIN 2.5V TO 5.5V CIN RPGD
L1 VIN LTC3606B RUN PGOOD RLIM GND RLIM CLIM VFB R2 R1 COUT SW CF VOUT
3606B F03a
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 3a. LTC3606B Layout Diagram (See Board Layout Checklist)
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LTC3606B APPLICATIONS INFORMATION
GND VIA TO VOUT SENSE VIN
GND RLIM GND SW
VFB RUN
PGOOD
VIN
SW GND VOUT
Figure 3b. LTC3606B Suggested Layout
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LTC3606B APPLICATIONS INFORMATION
Design Example As a design example, consider using the LTC3606B in a USB-GSM application, with VIN = 5V, IINMAX = 500mA, with the output charging a SuperCap of 4.4mF. The load requires 800mA in active mode and 1mA in standby mode. The output voltage VOUT = 3.4V. First, calculate the inductor value for about 40% ripple current (320mA in this example) at maximum VIN. Using a derivation of Equation (1): L1= 3.4V 3.4V *1 =1.51H 2.25MHz * (320mA) 5V of CIN = 10F should suffice, if the source impedance is very low. The feedback resistors program the output voltage. To maintain high efficiency at light loads, the current in these resistors should be kept small. Choosing 10A with the 0.6V feedback voltage makes R1~60k. A close standard 1% resistor is 59k. Using Equation (2). R2 = VOUT 1 * R1= 276k, 280k for 1% 0.6
For the inductor, use the closest standard value of 1.5H. The 4.4mF supercaps are used to deliver the required 2A pulses to power the RF power amplifiers, while the LTC3606B recharges the supercap after the pulse ends, see Figure 4c. As for the input capacitor, a typical value
A feedforward capacitor is not used since the 4.4mF SuperCap will inhibit any fast output voltage transients. Figure 4 shows the complete schematic for this example, along with the efficiency curve and transient response. Input current limit is set at 475mA average current, RLIM . = 116k, CLIM = 2200pF See Programming Input Current Limit section for selecting RLIM and Selection of CLIM Capacitance section for CLIM.
L1 1.5H
VIN USB INPUT 5V CIN 10F RPGD 499k
VIN LTC3606B RUN PGOOD RLIM
SW
VOUT 3.4V AT 800mA
R2 280k VFB R1 59k
+
PGOOD CLIM 2200pF RLIM 116k
COUT 2.2mF 2 SuperCap
GND
ILIM = 475mA CIN: AVX 08056D106KAT2A COUT: VISHAY 592D228X96R3X2T20H L1: COILCRAFT LPS4012-152ML
3606B F04
Figure 4a. Design Example Circuit
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LTC3606B APPLICATIONS INFORMATION
100 90 80 EFFICIENCY (%) 70 60 50 40 30 20 10 0 0.0001 VIN = 3.6V VIN = 4.2V VIN = 5V 0.001 0.01 0.1 OUTPUT CURRENT (A) 1
3606B F04b
VOUT = 3.4V
10
1 POWER LOSS (W)
0.1
0.01
0.001
Figure 4b. Efficiency vs Output Current
VOUT 200mV/DIV VIN 1V/DIV AC-COUPLED IOUT 500mA/DIV
IIN 500mA/DIV 1ms/DIV VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 2200pF , ILOAD = 0A TO 2A, COUT = 4.4mF VOUT = 3.4V ILIM = 475mA
3606B F04c
Figure 4c. Transient Response
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LTC3606B PACKAGE DESCRIPTION
DD Package 8-Lead Plastic DFN (3mm x 3mm)
(Reference LTC DWG # 05-08-1698)
0.70 0.05
3.5 0.05 1.65 0.05 2.10 0.05 (2 SIDES) PACKAGE OUTLINE 0.25 0.05 0.50 BSC 2.38 0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED R = 0.125 TYP 5 0.40 8 0.10
3.00 0.10 (4 SIDES) PIN 1 TOP MARK (NOTE 6)
1.65 0.10 (2 SIDES)
(DD8) DFN 0509 REV C
0.200 REF
0.75 0.05
0.25
4 0.05 2.38 0.10
1 0.50 BSC
0.00 - 0.05
BOTTOM VIEW--EXPOSED PAD
NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE
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LTC3606B REVISION HISTORY
REV A DATE 3/10 DESCRIPTION Changes to Electrical Characteristics PAGE NUMBER 3
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3606B TYPICAL APPLICATIONS
800mA Buck Converter, ILIM = 500mA
VIN USB INPUT 5V CIN 10F PGOOD CLIM 1000pF RPGD 499k L1 1.5H VIN LTC3606B RUN PGOOD RLIM GND RLIM 110k VFB R2 1210k R1 255k SW VOUT 3.4V AT 800mA
+
COUT 2.2mF 2 SuperCap
CIN: AVX 08056D106KAT2A COUT: VISHAY 592D228X96R3X2T20H
L1: COILCRAFT LPS4012-152ML
3606B TA02
800mA Buck Converter, ILIM = 475mA or Disabled
VIN USB INPUT 5V CIN 10F PGOOD ILIM DISABLE RLIM 116k RPGD 499k L1 1.5H VIN LTC3606B RUN PGOOD RLIM GND CLIM 2200pF VFB SW R2 1210k R1 255k VOUT 3.4V AT 800mA
+
COUT 2.2mF x2 SuperCap
CIN: AVX 08056D106KAT2A COUT: VISHAY 592D228X96R3X2T20H
L1: COILCRAFT LPS4012-152ML
3606B TA03
RELATED PARTS
PART NUMBER LTC3619/LTC3619B LTC3127 LTC3125 LTC3417A/ LTC3417A-2 LTC3407A/ LTC3407A-2 LTC3548/LTC3548-1/ LTC3548-2 LTC3546 DESCRIPTION Dual 400mA and 800mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converter 1.2A IOUT, 1.6MHz, Synchronous Buck-Boost DC/DC Converter with Adjustable Input Current Limit 1.2A IOUT, 1.6MHz, Synchronous Boost DC/DC Converter with Adjustable Input Current Limit Dual 1.5A/1A, 4MHz, Synchronous Step-Down DC/DC Converter Dual 600mA/600mA, 1.5MHz, Synchronous Step-Down DC/DC Converter Dual 400mA and 800mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converter Dual 3A/1A, 4MHz, Synchronous Step-Down DC/DC Converter COMMENTS 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 50A, ISD < 1A, MS10E, 3mm x 3mm DFN-10 94% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MAX) = 5.25V, IQ = 18A, ISD < 1A, 3mm x 3mm DFN-MSOP10E 94% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MAX) = 5.25V, IQ = 15A, ISD < 1A, 2mm x 3mm DFN-8 95% Efficiency, VIN(MIN) = 2.3V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V, IQ = 125A, ISD = <1A, TSSOP-16E, 3mm x 5mm DFN-16 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD = <1A, MS10E, 3mm x 3mm DFN-10 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 40A, ISD = <1A, MS10E, 3mm x 3mm DFN-10 95% Efficiency, VIN(MIN) = 2.3V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 160A, ISD = <1A, 4mm x 5mm QFN-28
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20 Linear Technology Corporation
(408) 432-1900 FAX: (408) 434-0507
LT 0310 REV A * PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2009

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